Specifically, power supplies operating at low power levels, typically less than 150 W, will be considered.
One aim is to minimize the size and weight of the power supplies.
“Flyback” or “forward” power supplies are low-power switch mode power supplies employed frequently, particularly because they are simple to control. The flyback design is a very interesting case because of its reduced size arising from the fact that it needs only one magnetic element to achieve the power conversion.
It will be recalled that in switch mode power supplies the DC voltage is chopped by a switch that is switching on and off at a frequency called the switching frequency.
A flyback power supply configuration and a forward power supply configuration will now be described; these are examples chosen from various known configurations.
A flyback power supply, a circuit diagram of which is shown in FIG. 1a), is an energy storage switch mode power supply.
It comprises a primary circuit P consisting of, in series, a voltage source Vin, a switch M, for example a MOS transistor and an inductor Lp made up of a winding of Np turns, and a secondary circuit S consisting of, in series, an inductor Ls made up of a winding of Ns turns, magnetically coupled to Lp, a capacitor Cout connected to a load represented here by a resistor Rload and a rectifier D, for example a diode.
For each of the windings of Lp and Ls, the phase φ, corresponding to the direction of the winding, is identified by a circle. In the example shown, the first and second windings have the same phase.
The coupling circuit consisting of the primary inductor Lp and the secondary inductor Ls is denoted by the transformer T.
The current flowing through the primary circuit is ip, and the voltages across the terminals of the primary circuit and across the switch are Vin and VM respectively. The current flowing through the secondary circuit is is, and the voltages across the terminals of the secondary circuit and across the diode are Vout and VD respectively.
In this “flyback” power supply design, current does not flow through both windings simultaneously. The operation of this power supply, called an “inductive storage” supply, is based on energy transfer cycles made up of a magnetic energy storage phase in the inductive element of the primary circuit (in this case Lp), followed by a phase for transferring this stored energy to a secondary source via the secondary circuit.
The various operating phases of this power supply will now be described, with reference to FIGS. 1b) and 1c).
Let us first recall a basic principle that underlies some of the explanations to follow: it is impossible to force a voltage discontinuity across the terminals of a capacitor and a current discontinuity in an inductor.
When the switch M is closed (FIG. 1b), i.e. during Ton, the energy is stored in the inductor Lp; the diode does not conduct since the voltage VD across its terminals is negative and therefore the current is is zero.
When the switch M is open, i.e. during Tswt-Ton, where Tswt is the switching period, the current ip is zero (FIG. 1c). The continuity of the magnetic energy leads to the transfer of the energy stored previously in the inductor Lp to the inductor Ls and also results in the diode D switching to its conducting state: D demagnetizes the transformer T. This phase ends if the current in the diode D falls to zero or if the end of the switching period is reached.
FIGS. 1d) and 1e) show the waveforms in continuous mode, in which the current is does not fall to zero at the end of the conducting phase of the secondary-circuit diode D. To simplify the description, it is assumed that the current ip changes instantaneously from its maximum value to zero.
The voltage VLp across the terminals of the inductor Lp, represented in FIG. 1d), varies as a function of time between a maximum value of Vin and a minimum value of −Vout×Np/Ns.
The current ip, represented in FIG. 1e), varies as a function of time between a maximum value of iMax and zero; the current is varies as a function of time between zero and a maximum value of iMax×Ns/Np.
A forward power supply, a circuit diagram of which is shown in FIG. 2a), is a switch mode power supply that directly transfers energy.
It comprises a primary circuit P consisting of, in series, a voltage source Vin, a switch M, for example a MOS transistor and an inductor Lp made up of a winding of Np turns, and, in parallel with the inductor Lp and the switch M, a demagnetizing circuit for demagnetizing the transformer which circuit may be a diode Ddem placed in series with an inductor Ldem, magnetically coupled to Lp, made up of a winding of Ndem turns. The diode Ddem and the inductor Ldem may be replaced by other components.
The secondary circuit S consists of, in series, an inductor Ls made up of a winding of Ns turns, magnetically coupled to Lp, a capacitor Cout connected to a load represented here by a resistor Rload, an inductor L, a first rectifier D1, for example a diode, and, in parallel with the inductor Ls and the rectifier D1, a second rectifier D2 which may also be a diode.
The phase φ of each of the windings of Lp, Ldem and Ls is identified by a circle. In the example given, the windings of Lp and Ls have the same phase, opposite to that of the winding of Ldem.
As in the previous case, the coupling circuit consisting of the primary inductor Lp, the secondary inductor Ls and the inductor Ldem is denoted by the transformer T.
The current flowing through the primary circuit is ip, and the voltages across the terminals of the primary circuit and across the switch are Vin and VM respectively. The current flowing through the secondary circuit is is, and the voltages across the terminals of the secondary circuit and across the diode D1 are Vout and VD1 respectively.
In this “forward” power supply design, both windings operate simultaneously; there is a direct transfer of energy between the inductors Lp and Ls.
The various operating phases of this power supply will now be described, with reference to FIGS. 2b), 2c) and 2d).
When the switch M is closed (FIG. 2b), i.e. during Ton, some of the energy is stored in the inductor Lp (this energy is a “parasitic” quantity and therefore much less than the energy of the direct transfer) and the remaining energy is directly transferred between the inductors Lp and Ls, and the diode D1 conducts; a current is flows in the secondary circuit; the diodes D2 and Ddem become nonconducting since the voltages across their terminals are negative.
When the switch M is open (FIG. 2c), the diode D1 becomes nonconducting, and the diodes D2 and Ddem switch to the conducting state. In accordance with the basic principle stated earlier, the diode D2, called a freewheeling diode, provides continuity of the current is in the inductor L and the diode Ddem provides continuity of the magnetic energy stored in the inductor Lp during the previous phase (i.e. during Ton) by transferring this stored energy to Vin over a time given by Ton×Ndem/Np: Ddem demagnetizes the transformer T.
At the end of the demagnetizing phase (FIG. 2d), i.e. during Tswt−Ton×(1+Ndem/Np), Ddem becomes nonconducting; D2 remains conducting. This is the freewheeling phase.
FIGS. 2e) and 2f) show the waveforms. To simplify the description, it is assumed that the current ip changes instantaneously from its maximum value to zero.
The voltage VLp across the terminals of the inductor Lp, represented in FIG. 2e), varies as a function of time between Vin and −Vin×Np/Ndem.
The current ip, represented in FIG. 2f), varies as a function of time between a maximum value of iMaxp and zero; the current is varies as a function of time between a maximum value of iMaxs and a minimum value of imin.
From now on, it will be generally assumed that a primary circuit P includes at least one switch M placed in series with a voltage source Vin and a first inductor Lp, that a secondary circuit includes at least one rectifier D placed in series with a second inductor Ls and a capacitor Cout connected to a load, and that the primary and secondary circuits are coupled by a coupling circuit including at least the primary inductor Lp and the secondary inductor Ls magnetically coupled to each other.
One aim is to further reduce the size and weight of these power supplies.
In order to be able to use small components while achieving the same energy conversion possibilities in terms of the power available at the output, the switching frequency must be increased. This has the drawbacks of increasing losses in the transformer and switch-related losses in the other components, which in turn reduces the overall efficiency and therefore raises the temperature and reduces reliability.
High-frequency imperfections in the transformer are conventionally modeled by a leakage inductance Lf in series with the inductor Lp, as shown in FIG. 3a) for a flyback power supply and in FIG. 3b) for a forward power supply.
In the case of a flyback power supply operating in discontinuous mode, in which the current is falls to zero at the end of the conducting phase of the secondary-circuit diode D, the voltage across the terminals of the switch M when it opens can be given approximately by the following formula:       V    M    =            V      in        +                  Np        Ns            ×              V        out              +                  L        f            ×              ⅆ                  ⅆ          t                    ⁢                          ⁢              I        p            
When the switch opens, it is assumed that the current decreases linearly from its maximum value to zero over a time Tfall which is the closed/open switching time of the switch. Therefore, upon opening of the switch M, and with Ton being the time over which the switch M is closed:       V    M    =            V      in        +                  Np        Ns            ×              V        out              +                            L          f                          L          p                    ×              V        in            ×                        T          on                          T          fall                    
Hence, the leakage inductance results in a term representing an overvoltage across the terminals of the switch, in the form:                     L        f                    L        p              ×          V      in        ×                  T        on                    T        fall              ,and the power Pf due to the leakage inductance is:             P      f        =                  1        2            ×                                    V            in            2                    ×                      T            on            2                                    L          p          2                    ×              1                  T          swt                    ×              L        f              ,where Tswt is the switching period. The energy stored in the leakage inductance is in general dissipated during the switching phases.
Furthermore, the switching-related losses upon opening of the switch are proportional to Tfall.
Therefore, reducing the switching time Tfall reduces the switching-related losses but increases the term representing the overvoltage across the terminals of the switch.
For example, an opening time Tfall 100 times lower than the closure time Ton, and a leakage inductance Lf of about 1% of Lp, results in an overvoltage upon the switching action equal to the power supply voltage Vin. The consequences of this would be disastrous as regards the voltage dimensioning of the switch M, in this case the transistor M which must be a high voltage range transistor and therefore more expensive and less effective.
In the case of a forward power supply, other equations are derived but the same observations are made on interpreting them.
There are several types of circuits for countering the effect of the leakage inductance.
Dissipative RCD (i.e. Resistor, Capacitor, Diode) circuits are very effective in limiting overvoltages but they dissipate all the energy stored in the leakage inductance resulting in a reduction in overall efficiency.
FIG. 4 shows an example of a flyback power supply employing an RCD circuit. The capacitor C limits the term representing the overvoltage upon opening of the switch M; the resistor R discharges the voltage across the terminals of C and thus dissipates the energy stored in the leakage inductance.
Snubber circuits are often employed to reduce the overvoltages across the terminals of the switch M.
FIG. 5 shows an example of a flyback power supply employing a snubber circuit that dissipates very little energy. As in the previous case, the capacitor limits the overvoltages across the terminals of the switch M. To recover the energy stored in C, an oscillating circuit based on L and C inverts the voltage across the terminals of C. In practice, losses in diodes D1 and D2 and in the inductor L limit the portion of energy recovered by the circuit. Furthermore, the oscillations of the LC circuit must be damped, which also reduces the efficiency.
Lastly, such a circuit is more complex and therefore less reliable, and the efficiency of the power supply would have improved only slightly.